High probability of intercept radar detector

ABSTRACT

A detector for detecting continuous wave police radar that includes an antenna configured to receive an input signal, a diplexer in communication with the antenna to separate the input signal into a high-band signal and a low-band signal, a local oscillator configured to sweep through a range of frequencies to produce F LO , and a frequency multiplier to generate a first mixing signal that is an integer multiple of F LO . The detector also includes a high-band intermediate-frequency signal and a low-band intermediate-frequency signal with a switch configured to select one of them as an output intermediate-frequency signal. A second-stage mixes the output intermediate-frequency signal with F LO  to generate an output signal, and a determination is made whether the input signal includes a police radar signal.

RELATED APPLICATIONS

The present application is a continuation-in-part of, and claims benefit from, U.S. application Ser. No. 13/834,905, filed Mar. 15, 2013, entitled HIGH PROBABILITY OF INTERCEPT RADAR DETECTOR, the contents of which are incorporated herein by reference in their entirety.

FIELD OF THE INVENTION

The present invention relates generally to police radar detectors used in motor vehicles and, more particularly, to decreasing detector sweep time when detecting radar signals.

BACKGROUND

Conventional radar detectors almost universally employ scanning super-heterodyne receiver architectures. To achieve good sensitivity with lower cost, conventional radar detectors tend to sweep relatively slowly, often requiring several tenths of a second to sweep a covered spectrum. As a result, some radar gun manufacturers have developed radar gun designs that transmit very brief pulses as a technique for avoiding detection. The brief transmission is of a relatively short duration and may be conveniently referred to as a “POP transmission” which is a phrase coined and trademarked by MPH Industries. A detector with sweep periods lasting several tenths of a second is likely to entirely miss a radar gun transmission that lasts only in the neighborhood of no more than several tens of milliseconds. One approach for a detector design may involve sweeping the spectrum much faster to try to intercept these brief transmissions. In this way, the detector will tune through the transmission frequency during the interval that the radar signal is actually present. However, this approach will greatly increase the required bandwidth of the detector's receiver. Because the received signal power remains unchanged in the increased bandwidth, the signal-to-noise ratio degrades and a commensurate decrease in threshold sensitivity occurs.

Thus, there remains a need for another approach to reliably detecting these brief radar gun transmissions that avoids undue loss of the signal-to-noise ratio.

SUMMARY

One aspect of the present invention relates to a detector for detecting continuous wave police radar that includes a) an antenna configured to receive an input signal, the input signal comprising a continuous wave emission within at least one radar band, b) a diplexer in communication with the antenna to separate the input signal into a high-band signal having a first range of frequencies and a low-band signal having a second range of frequencies, the second range of frequencies less than the first range of frequencies, c) a local oscillator configured to sweep through a range of frequencies from a first frequency to a second frequency in a predetermined time period to produce a signal F_(LO), and d) a frequency multiplier in communication with the local oscillator signal to generate a first mixing signal with a frequency that is positive, integer multiple of F_(LO). The detector also includes a high-band first-stage mixer configured to mix the first mixing signal with the high-band signal to produce a high-band intermediate-frequency signal; a low-band first-stage mixer configured to mix the signal F_(LO) with the low-band signal to produce a low-band intermediate-frequency signal; and a switch coupled with the high-band first-stage mixer and the low-band first-stage mixer, configured to select one of the high-band intermediate frequency signal and the low-band-frequency signal as an output intermediate-frequency signal. Next, the detector includes a second-stage mixer in communication with the switch configured to mix the output intermediate-frequency signal with the signal F_(LO) to generate an output signal F_(O); and a signal analyzer configured to determine whether the input signal received by the antenna includes a police radar signal based on the output signal F_(O).

Still a further aspect of the present invention relates to a method for detecting continuous wave police radar that includes receiving, by an antenna, an input signal, the input signal comprising a continuous wave emission within at least one radar band and separating, by a diplexer in communication with the antenna, the input signal into a high-band signal having a first range of frequencies and a low-band signal having a second range of frequencies, the second range of frequencies less than the first range of frequencies. The method also includes sweeping, by a local oscillator, through a range of frequencies from a first frequency to a second frequency in a predetermined time period to produce a signal F_(LO), and generating, by a frequency multiplier in communication with the local oscillator signal to generate a first mixing signal with a frequency that is positive, integer multiple of F_(LO). Next, the method continues with mixing, by a high-band first-stage mixer, the first mixing signal with the high-band signal to produce a high-band intermediate-frequency signal; and mixing, by a low-band first-stage mixer, the signal F_(LO) with the low-band signal to produce a low-band intermediate-frequency signal; so that a switch, coupled with the high-band first-stage mixer and the low-band first-stage mixer, can select one of the high-band intermediate frequency signal and the low-band-frequency signal as an output intermediate-frequency signal. The method concludes with mixing, by a second-stage mixer in communication with the switch, the output intermediate-frequency signal with the signal F_(LO) to generate an output signal F_(O); and determining, by a signal analyzer, whether the input signal received by the antenna includes a police radar signal based on the output signal F_(O).

BRIEF DESCRIPTION OF THE DRAWINGS

While the specification concludes with claims particularly pointing out and distinctly claiming the present invention, it is believed that the present invention will be better understood from the following description in conjunction with the accompanying Figures, in which like reference numerals identify like elements, and wherein:

FIG. 1A depicts a block-level diagram of a radar search receiver with chirp compression.

FIG. 1B depicts a block-level diagram of portions of a radar signal detector in accordance with the principles of the present invention.

FIG. 2A depicts major components of the Ka-band signal path from the front antenna of the radar signal detector of FIG. 1B.

FIG. 2B depicts major components of the K-band signal path from the front antenna of the radar signal detector of FIG. 1B.

FIG. 2C depicts major components of the Ku-band signal path from the front antenna of the radar signal detector of FIG. 1B.

FIG. 2D depicts major components of the X-band signal path from the front antenna of the radar signal detector of FIG. 1B.

FIG. 3A depicts an exemplary continuous wave radar signal such as those emitted from a police radar gun.

FIG. 3B and FIG. 3C depict, respectively, a linear down-chirp signal and a linear up-chirp signal in accordance with the principles of the present invention.

FIG. 3D depicts a compressed pulse resulting from propagating a linear chirp signal through a matched filter in accordance with the principles of the present invention.

FIG. 4A depicts a conceptual graph of an up-chirp.

FIG. 4B depicts a conceptual graph of a down-chirp.

FIG. 4C depicts a graph of the frequency-to-time ratio of the swept local oscillator when sweeping through the Ka-band in accordance with the principles of the present invention.

FIG. 5 depicts the general structure of the local oscillator waveform for a particular radar band sweep in accordance with the principles of the present invention.

FIG. 6 depicts a flowchart of an exemplary sweep process in accordance with the principles of the present invention.

FIG. 7 depicts representative signal shapes for a detected RF signal and an undesired image signal.

FIG. 8A depicts a flowchart of an exemplary process to implementing a fixed detection window in accordance with the principles of the present invention.

FIG. 8B depicts a block level diagram of utilizing a dynamically adjusted noise floor in accordance with the principles of the present invention.

FIG. 9 depicts what the activation of a radar gun may look like when detected in accordance with the principles of the present invention.

FIG. 10 depicts an alternative to the block-level diagram of FIG. 1B of portions of a radar signal detector in accordance with the principles of the present invention.

FIG. 11A depicts major components of the Ka-band signal path from the front antenna of the radar signal detector of FIG. 10.

FIG. 11B depicts major components of the K-band signal path from the front antenna of the radar signal detector of FIG. 10.

FIG. 11C depicts major components of the Ku-band signal path from the front antenna of the radar signal detector of FIG. 10.

FIG. 11D depicts major components of the X-band signal path from the front antenna of the radar signal detector of FIG. 10.

DETAILED DESCRIPTION

In the following detailed description of the preferred embodiment, reference is made to the accompanying drawings that form a part hereof, and in which is shown by way of illustration, and not by way of limitation, specific embodiments in which the invention may be practiced. It is to be understood that other embodiments may be utilized and that changes may be made without departing from the spirit and scope of the present invention.

An inherent property of a matched filter is that it has an impulse response that is a time-reversed replica of the desired signal (plus a time delay required for causality). In a wideband search application using a compressive receiver design similar to one as described below, one example desired signal structure is a linear chirp. Compressive receiver design may reasonably begin with selection of a compressive filter since it is one of the most challenging receiver components. One example compressive filter type is a dispersive delay line (DDL) surface acoustic wave (SAW) filter which is generally fabricated to have an impulse response that is a down-chirp rather than an up-chirp because the down-chirp devices tend to deliver better performance (e.g. lower loss) than do up-chirp devices. Assuming selection of a DDL SAW filter having the favored down-chirp impulse response, the matched-signal input would be an up-chirp and accordingly the receiver design would include an oscillator swept so as to deliver up-chirps of the proper rate to an input of the DDL SAW filter.

An example super-heterodyne receiver architecture described below linearly sweeps the local oscillator (LO) to down-convert received continuous wave radio frequency (CW RF) signals into linear up-chirps at the receiver's intermediate frequency (IF) output (i.e. the matched filter input). In one example embodiment described herein, the receiver's RF frequency conversion conforms to the formula: IF=RF−N*LO where N=2, 3, 5, 7 to sweep X, Ku, K, and Ka, respectively. Therefore, to deliver signal up-chirps at the IF, the LO generates down-chirps. One of ordinary skill will recognize opposite behavior can be designed and accomplished if the compressive filter conversely has an input response that happens to be an up-chirp.

By way of example, a receiver design may include a DDL filter that has a matched signal input chirp rate of +3.5×10¹² Hz/s, with center frequency of 280 MHz. In accordance with matched-filter theory, the filters impulse response is a burst down-chirp centered at 280 MHz, with a chirp rate of −3.5×10¹² Hz/s. The duration of the impulse-response burst is established by specifying the required dispersive time-delay length of the filter. This dispersive delay, or dispersive time-delay length, is the differential delay between the travel times of the frequency extremes of the DDL filter. The acoustic velocity on a surface of a SAW is essentially constant for all frequencies; however, the effective path length to an array of resonators metalized on the surface varies with frequency. Thus, constant velocity over varying lengths gives rise to varying delay times for different frequencies. Subsequent description refers to “delay” for convenience, but it should be understood that dispersive time-delay length is the salient parameter under discussion. The bandwidth of the impulse burst is simply the dispersive time-delay length times the chirp rate and this bandwidth can also be characterized as the signal bandwidth of the filter. The dispersive time-delay length is one performance parameter of the filter that governs the amount of signal energy that can be captured by the filter and, thus, governs the filter's performance advantage. In principle, a receiver's performance improves with an increase in the time-delay length of the filter. For example, a filter having a dispersive time-delay length between about 2 ρs to about 20 μs provides beneficial results in the frequency ranges typically encountered with police-band radar signals. In particular, an example dispersive time-delay length of 4 μs is used in an example design described below to help facilitate understanding of the principles of the present invention.

FIG. 1A depicts a block-level diagram of a radar search receiver with chirp compression. For clarity, intermediate components such as filters and amplifiers are omitted from the idealized diagram of FIG. 1A. A continuous wave radar signal is received by an antenna 52 and is fed, as signal “RF” to a mixer 54. An example of such a signal is depicted in FIG. 3A which shows a fixed frequency signal pulse 300 over a period of time.

Another input to the mixer 54 is a swept local oscillator signal or “LO” signal generated by a linear sweep generator 60. The linear sweep generator 60 generates a signal whose frequency changes in a linear fashion over time. If the frequency of the generated signal increases as time progresses, then this is conveniently referred to as an “up-chirp”. If the generated signal decreases in frequency as time progresses, then this is conveniently referred to as a “down-chirp”. As used herein, a down-chirp signal may be described as having a chirp rate of X Hz/s because the term “down-chirp” provides an indication that the chirp rate frequency is decreasing at a rate having a magnitude of X Hz/s. Equivalently, for clarity, a down-chirp signal, even when identified as a “down-chirp”, can be described as having a chirp rate of −X Hz/s as a way to emphasize its decreasing rate. FIG. 3B depicts a down-chirp sweep signal 302 that may be produced by the linear sweep generator 60 and FIG. 3C conversely depicts an example up-chirp sweep signal 303. FIGS. 4A and 4B depict respective chirp signals as a graph of frequency versus time. The signal of FIG. 4A is an up-chirp signal with a positive chirp rate 402 that produces an increasing frequency as time progresses; while the signal of FIG. 4B is a down-chirp signal with a negative chirp rate 404 that produces a decreasing frequency as time progresses.

When the RF signal and the LO signal are combined at the mixer 54, then the RF signal is frequency shifted by the instantaneous frequency of the LO signal. In particular, mixing an RF signal having a frequency f_(RF) and a LO signal having a frequency f_(LO) produces two different signals at a respective intermediate frequency. One of the produced signals will have an intermediate frequency of f_(RF)+f_(LO) and the other produced signal will have an intermediate frequency of f_(RF)−f_(LO). In FIG. 1A, the “minus” sign next to mixer's input for the LO signal indicates that the filter of FIG. 1A happens to pass f_(IF)=f_(RF)−f_(LO). It should also be noted that f_(IF) is proportional to −f_(LO); as a result, a down-chirping LO will be reflected as an up-chirping signal in the IF chirp output.

When a continuous wave signal such as that in FIG. 3A is mixed with a down chirp signal as shown in FIG. 3B, the resulting signal, “IF CHIRP”, from the mixer 54 will be an up-chirp signal similar to the signal 303 of FIG. 3C. A filter 56 is selected that is a matched filter with respect to the shape of the IF CHIRP signal. As described earlier, a matched filter is one that has an impulse response with exactly the same shape as its input signal but reversed in time with an added delay related to the duration of the input signal.

Chirp signals are characterized by a chirp rate, as shown if FIGS. 4A and 4B. For example, the linear sweep generator 60 may produce a signal whose frequency linearly changes at a rate of −3.5 MHz/μs. Equivalently, this signal would have a chirp rate of −3.5×10¹² Hz/s. The inverting arrangement of the mixer 54 results in the signal at the IF output (i.e., the desired signal) having a chirp rate of 3.5×10¹² Hz/s. As for matched filter selection, the matched filter impulse response for a chirp signal is itself a chirp signal having the same magnitude but with an opposite sign than that of the desired signal. Accordingly, for the example signal just described, the matched filter would have an impulse response chirp rate of −3.5×10¹² Hz/s; alternately stated, the filter's matched signal is an up-chirp at 3.5×10¹² Hz/s, precisely what is delivered by the sweep generator/mixer combination. This particular chirp rate is provided merely by way of example and other chirp rates having different magnitudes, either greater than or less than 3.5×10¹² Hz/s, are contemplated as well.

The result of the up-chirp linear FM chirp signal 303 propagating through the matched chirp compression filter 56 will be a compressed pulse similar in shape to the signal 304 of FIG. 3D. In particular, the signal 304 is a sinc waveform having a peak occurring at a particular time T 306 that is based on the frequency of the RF signal. Accordingly, a detector 58 can detect an occurrence of a peak in the signal 304 and convey this occurrence to control and analyzer circuitry 62. The control and analyzer circuitry 62 can then determine if the detected signal is likely a received radar transmission and also determine the frequency of that radar transmission. The control and analyzer circuitry 62 can also be coupled with the linear sweep generator 60 to control the LO waveform that is being provided to the mixer 54.

Additionally, a user interface 63 can include audible or visual indicators that are representative of the signals being received and detected as well as allow a user to select different operating characteristics of the device of FIG. 1A.

Applying these principles to the detection of radar gun transmissions, there is a continuous wave signal at an unknown frequency somewhere in the radar bands, a spectrum that can total about 3 GHz. A super-heterodyne receiver remains a viable technique for scanning through this wide spectrum. As a result, the received RF signal will be down-converted to an intermediate frequency signal that is a linear chirp. Thus, a matched filter will also have an impulse response that is a linear chirp. Such a matched filter can be realized as a dispersive delay line acting as a receiving filter. In this configuration of a receiving filter, as the intermediate frequency chirp propagates through the dispersive delay line, the delay line accumulates the energy in the matching chirp during the several microseconds the signal is present within the delay line. The delay line acts to compress the matched chirp and delivers the stored energy as a narrow output pulse only a few tens of nanoseconds wide.

A matched filter which captures as much signal energy as possible provides the best sensitivity. However, in practice, a radar detector is desired that will intercept very brief radar pulses and, thus, a sweep chirp rate should be fast enough to cover the entire spectrum in less than the length of a POP transmission. The length of a POP transmission may, for example, be about 16 ms. In this example, the entire sweep cycle should be completed in less than 16 ms to ensure interception of the limited duration POP transmission. This principle should govern irrespectively of the number of antennae employed or the number of radar bands swept. To elaborate, for reasons of economy, it may be desirable to share the DDL filter and other receiver circuitry sequentially amongst multiple bands and/or antennae. Receiver sensitivity might have to be compromised commensurately, but completing the sweep cycle often enough to ensure POP signal interception should be the guiding constraint. These considerations will be evident to those having ordinary skill. Some circumstances may motivate sweeping faster than 16 ms. If both a front and rear antenna are considered, in a sequential manner, then the sweep should be able to be performed in less than 8 ms. If multiple sweeps are desired in order to identify spurious signals or false alarms, then even faster sweep rates are beneficial. FIG. 1B depicts a block-level diagram of portions of a radar signal detector in accordance with the principles of the present invention. There are three particular radar bands that it is typically desirable to sweep through in order to detect radar gun transmissions. The X-band center is nominally at 10.525 GHz but a desirable sweep will include the range of 10.49-10.56 GHz; The K-band center is nominally at 24.150 GHz but a desirable sweep will include the range of 24.035-24.265 GHz; and the Ka-band center is nominally at 34.7 GHz but a desirable sweep will include the range of 33.35-36.08 GHz. A fourth band may also be swept: the Ku-band center is nominally at 13.45 GHz but a desirable sweep will include the range of 13.38-13.52 GHz. These frequencies encompass the FCC allocated police radar bands plus modest “over-scan” for safety.

The diagram of FIG. 1B shows different switches (e.g., 112A, 112B, 118, 126, 158) that define the signal path through the radar signal detector at a particular time. These switches may be implemented in a variety of different ways without departing from the scope of the present invention. For example, one or more of the switches may be a transistor having its gate voltage set so as to be transmissive or non-transmissive in a given direction or various switches can be functionally achieved by activating a bias on amplifiers in a desired path and deactivating amplifiers in unwanted paths. The microprocessor 180 and/or field programmable gate array (FPGA) 176, or similar control circuitry, can control the switches so as to select a current signal path and, thereby, select a particular band of radar signals of interest at a particular time.

When considering signals arriving at a front antenna 102, the switches 112A and 112B will select the appropriate signal path. Thus, an RF signal at the front antenna 102 will pass to the diplexer 104 and be split into K-band and Ka-band frequencies on one path and X-band and Ku-band frequencies on a different path. The K-band and Ka-band frequencies can be amplified by an amplifier 106 and mixed with a local oscillator based signal at a mixer M1F 108. The output of the mixer 108 can be amplified by amplifiers 110, 116, and 128 before propagating to another diplexer 144. At the diplexer 144, the K-band signals are separated from the Ka-band signals. K-band signals pass through the switch 158 and are amplified by an amplifier 160 before being mixed, at a mixer M4 162, with a local oscillator signal from the voltage controlled oscillator 140. The output F_(O) 164 of the mixer M4 162 is amplified by an amplifier 168 and propagated to a dispersive delay line filter 170. One of ordinary skill will recognize that the different bands can be combined and separated in ways that are different than those of FIG. 1B. For example, the K-band signals could be handled by a separate signal path while the Ka and X-band signals can be in a combined path from which they are alternately selectable.

At the diplexer 144, the desired band of signals could have been in the Ka-band which would have resulted in Ka-band signals being propagated from the diplexer 144 to a band-pass filter 146 and mixed, at a mixer M3 148, with a local oscillator based signal. The output from the mixer M3 148 can be filtered with another band-pass filter 150 and can then pass through the switch 158 to the amplifier 160 where it is then mixed in the mixer M4 162 with a local oscillator signal from the voltage controlled oscillator 140. The output F_(O) 164 from the mixer M4 162 can then be amplified using the amplifier 168 and propagated to the filter 170. At the mixer M4 162, an image signal 166 can be rejected by, for example, being shunted to ground or other known techniques.

Returning to diplexer 104, the other path is for received RF signals in either the X-band or the Ku-band. Signals in either of these bands may be amplified by an amplifier 107 and then a switch 118 can direct Ku-band signals directly to an amplifier 124 or can direct X-band signals to an amplifier 120. The output of the amplifier 120 is mixed at mixer M2 122 with the local oscillator signal from the voltage controlled oscillator 140 and propagated to the amplifier 124. A switch 126 can connect the output from the amplifier 124 to the amplifier 128 so that the diplexer 144 can direct the X-band and Ku-band signals to the band-pass filter 146 and from there to the mixer M3 148. At the mixer M3 148, the output from the filter 146 and a local oscillator based signal are mixed and propagated to the band-pass filter 150. The output from the filter 150 passes through the switch 158, the amplifier 160, the mixer M4 162, and the amplifier 168 to arrive at the filter 170.

When considering signals received by a rear antenna 138, similar signal paths are provided for each of the different bands of signals in the covered spectrum. However, these alternative signals paths utilize a diplexer 136, various amplifiers 132, 134, and 114, and a mixer M1R 130.

The circuit arrangement of FIG. 1B utilizes a single voltage controlled oscillator 140 to produce a local oscillator signal F_(LO). As is known in the art, a reference voltage generator 156, phase lock loop 152, and amplifier 154 work in conjunction with the voltage-controlled oscillator 140 to produce a linear sweep signal having desired characteristics for the band of signals being detected at a given moment. The signal F_(LO) may be doubled by a frequency doubler 142 and provided to various ones of the mixers (e.g., M1F, M1R, M3).

The microprocessor 180 and/or the FPGA 176 coordinates frequency control and receiver band switching. In particular, as mentioned, the frequency of the VCO 140 is controlled by the PLL 152 and reference generator 156. An example reference generator 156 may be a direct digital synthesizer (DDS) that generates a dynamically varying reference input to the PLL 152. The PLL 152, in turn, is used to frequency multiply the reference input to an appropriate frequency for F_(LO). Example devices may be Analog Devices AD9913 and ADF4106 for the DDS and PLL, respectively. Alternatively, Analog Devices ADF4158 may be used as the reference generator 156 that is capable of synthesizing sweep waveforms.

Ultimately, a received signal from any of the bands and from either the front or rear antenna will be propagated to the dispersive delay line filter 170. This filter can be implemented using a surface acoustic wave device or by digitizing the IF with an analog-to-digital converter and processing with an appropriately programmed digital signal processor. As will be discussed in more detail below, the filter 170 is a filter that is matched with the shape of the signal F_(O) produced as described above. The filter 170 produces an output that is detected by a demodulating log amplifier 172. The output of the log amplifier 172 is continuously digitized by a high-speed analog-to-digital converter 174 and input to the FPGA 176. The FPGA 176 digitally processes this input and determines if the output from the converter 174 exceeds a noise floor sufficiently to be deemed a real, detected signal.

An algorithm within the FPGA 176 may dynamically quantify the noise floor so that a detection threshold can adapt to fluctuations in noise power as the bands are being swept. When the FPGA 176 recognizes a signal, it stores (for example, in a FIFO buffer) the signal's peak amplitude as a measure of signal strength and attaches a timing “tag” that indicates the time when the peak was detected. If a side lobe 308 is falsely considered to be a detected pulse, then the pulse waveform 304 of FIG. 3D may inaccurately be detected as multiple closely-spaced signals. Accordingly, the FPGA 176 may use a detection window approach to avoid inaccurate results. The detection window is a period of time during which the FPGA 176 monitors the output from the converter 174 to search for the peak amplitude. In one embodiment, the FPGA 176 dynamically analyzes the converter output to determine when a pulse is over. A simpler approach is to have a fixed-size detection window wherein the window size is based on the expected pulse width at large signal strengths and a desired resolution of nearby signals.

The detector of FIG. 1B may also include a laser detector 184. In particular, the laser detector 184 can include a front sensor and a rear sensor for detecting police LIDAR signals. A power supply 182 can be included to ensure sufficient electrical power for all the components and a microprocessor 180 can be included to control the operation and timing of the appropriate elements of the detector of FIG. 1B.

In FIG. 1B, the radar signal detector was depicted in such a way as to highlight how a single local oscillator 140 is used to produce all of the mixer signals for all of the radar bands in the covered spectrum. This scheme offers a number of advantages with respect to hardware efficiency. Especially advantageous is the avoidance of multiple fixed LOs and their propensity to generate problematic mixing products that result in internal spurious signals. However, it should be obvious that numerous other receiver frequency schemes could be adopted without departing from the spirit of the invention.

To explain operation of the circuitry in FIG. 1B, FIGS. 2A-2D are provided below to isolate each signal path for a respective band of radar signals. In these figures, the various switches, amplifiers, and filters have been omitted for clarity. For each radar band, the VCO 140 produces an appropriately swept F_(LO) that is different from the F_(LO) for the other bands. As depicted in FIG. 2A, a particular spot F_(LO) frequency (e.g., 5.1029 GHz) is shown corresponding to a particular RF signal frequency (e.g., 36 GHz) within a particular radar band (e.g., Ka-band).

Within each of the signal paths depicted in FIGS. 2A-2D there is a respective set of components comprising various mixers and a frequency doubler arranged in a particular way. For each radar band signal path, the respective set of components act together, along with the VCO 140, to achieve a composite local oscillator that functions similar to the linear sweep generator 60 of FIG. 1A. Thus, for each radar band, as F_(LO) is controlled to sweep through its particular range of frequencies at a particular rate, the respective set of components act in concert with one another to function as a composite local oscillator that sweeps through a range of frequencies at a particular rate to produce an intermediate frequency (IF) signal, F_(O), as input to a matched filter.

FIG. 2A depicts major components of the Ka-band signal path from the front antenna 102 of the radar signal detector of FIG. 1B. The sweep output F_(LO) from the VCO 140 is doubled by a frequency doubler 142 and the second harmonic of this doubled frequency is mixed, at mixer M1F 108, with an incoming Ka-band 36 GHz signal. The first intermediate frequency is in the range of 15.5886 GHz and is mixed, at mixer M3 148, with the doubled F_(LO) to produce a second intermediate frequency signal in the range of 5.3829 GHz. This second intermediate frequency signal is mixed, at mixer M4 162, with F_(LO) to produce an output signal F_(O) having an intermediate frequency in the range of 280 MHz. This signal is then propagated to the matched filter 170.

As mentioned, the sweep for Ka-band signals may sweep from 33.35 GHz to 36.08 GHz. Using the signal path of FIG. 2A, this sweep can be accomplished when F_(LO) is swept from 4.7242 GHz to 5.1142 GHz.

FIG. 2B depicts major components of the K-band signal path from the front antenna 102 of the radar signal detector of FIG. 1B. The sweep output F_(LO) is doubled by the frequency doubler 142 and its second harmonic is mixed, at mixer M1F 108, with an incoming K-band 24.25 GHz signal. The output of the mixer M1F has an intermediate frequency of about 5.074 GHz and is mixed, at mixer M4 162, with F_(LO) to produce an output signal F_(O) having an intermediate frequency in the range of 280 MHz.

As mentioned, the sweep for K-band signals may sweep from 24.035 GHz to 24.265 GHz. Using the signal path of FIG. 2B, this sweep can be accomplished when F_(LO) is swept from 4.751 GHz to 4.797 GHz.

FIG. 2C depicts major components of the Ku-band signal path from the front antenna 102 of the radar signal detector of FIG. 1B. The sweep output F_(LO) is doubled by the frequency doubler 142 and is mixed, at mixer M3 148, with an incoming Ku-band 13.5 GHz signal. The output of the mixer M3 has an intermediate frequency of about 4.6866 GHz and is mixed, at mixer M4 162, with F_(LO) to produce an output signal F_(O) having an intermediate frequency in the range of 280 MHz.

As mentioned, the sweep for Ku-band signals may sweep from 13.38 GHz to 13.52 GHz. Using the signal path of FIG. 2C, this sweep can be accomplished when F_(LO) is swept from 4.3667 GHz to 4.4133 GHz.

FIG. 2D depicts major components of the X-band signal path from the front antenna 102 of the radar signal detector of FIG. 1B. The sweep output F_(LO) is additively mixed, at mixer M2 122, with an incoming X-band 10.55 GHz signal to produce a first intermediate frequency signal in the range of 15.685 GHz. The sweep output F_(LO) is also doubled by the frequency doubler 142 and is mixed, at mixer M3 148, with the first intermediate frequency signal. The output of the mixer M3 148 has an intermediate frequency of about 5.415 GHz and is mixed, at mixer M4 162, with F_(LO) to produce an output signal F_(O) having an intermediate frequency in the range of 280 MHz.

As mentioned, the sweep for X-band signals may sweep from 10.49 GHz to 10.56 GHz. Using the signal path of FIG. 2D, this sweep can be accomplished when F_(LO) is swept from 5.105 GHz to 5.14 GHz.

As mentioned above, FIGS. 2A-2D are provided to illustrate an isolated signal path for each respective band of radar signals. In these figures, the various switches, amplifiers, and filters have been omitted for clarity and each of the signal paths happen to relate to the front antenna 102 of FIG. 1B. Similar distinct signal path diagrams can be constructed as well that relate to the rear antenna 138. To do so, FIGS. 2A-2D would be modified to use mixers that are in the signal paths related to the rear antenna 138. For the Ku-band and X-band, no modifications are needed; the same mixers are used regardless of whether the signal path relates to the front antenna 102 or the rear antenna 138. For the K-band and Ka-band, the mixer M1F 108 would be replaced with mixer M1R 130.

The matched filter 170, as mentioned, can be a dispersive delay line configured as a receiver filter. In an embodiment, the filter 170 can have a matched signal that is an up-chirp with a chirp rate of 3.5 MHz/μs. Accordingly, the sweep rate of F_(LO) can be controlled so that an RF signal at the receiver input is down-converted to the IF and arrives with the matching 3.5 MHz/μs chirp rate. Thus, when the FPGA 176 determines the Ka-band using the front antenna 102 is the band of interest, it will control all the appropriate switches in FIG. 1B to define the Ka-band signal path and it will control the VCO 140 to produce a sweep signal F_(LO) that matches the filter 170.

Using FIG. 2A as an example, the frequency F_(LO) can be characterized as

$F_{LO} = \frac{{Ka} - F_{o}}{7}$ where Ka=36 GHz and F_(O)=280 MHz, for example. Rearranging the terms reveals that F_(O)=Ka−7F_(LO). Thus, any change in the local oscillator signal F_(LO) is effectively multiplied by “7” when producing the sweep rate of the composite local oscillator that results in the output chirp F_(O) at the IF frequency of 280 MHz. This can be visualized in FIG. 4C which depicts a graph of the frequency-to-time ratio of the swept local oscillator in accordance with the principles of the present invention. The line 406 represents the sweep local oscillator F_(O) which has a slope, or chirp rate, of Δf/Δt. However, because of the way various multiples of F_(LO) are mixed in the signal path of FIG. 2A, the effective chirp rate 408 of the composite local oscillator comprised of the set of components in the signal path of FIG. 2A is 7 times Δf/Δt. When the incoming 36 GHz signal is mixed with the composite local oscillator signal represented by 408 of FIG. 4C, a resulting F_(O) signal is generated as input to the filter 170 and will have a center frequency at 280 MHz and be an up-chirp with a chirp rate of 3.5 MHz/μs. Embodiments of the present invention also contemplate slower chirp rates such as, for example, 0.15 MHz/μs and 2.85 MHz/μs and even chirp rates above 3.5 MHz/μs.

Thus, for the Ka-band the sweep local oscillator signal F_(LO) is not simply swept at the down chirp rate of −3.5 MHz/μs. Instead, a sweep rate of

$\frac{{- 3.5}\mspace{14mu}{{MHz}/{\mu s}}}{7}$ is used to produce a signal F_(O) that is matched to the matching rate of the filter 170.

Accordingly, the F_(LO) signal for the Ka-band is controlled by the FPGA 176 to sweep from a starting frequency to a stopping frequency at a specific sweep rate. The starting and stopping frequencies are selected to tune through the entire Ka-band and the sweep rate is selected to match the impulse response of the filter 170.

As shown in FIG. 2B-2C, a similar relationship between F_(LO) and a received radar signal band can be determined for the other three bands such as, for example, by:

${{for}\mspace{14mu}{the}\mspace{14mu} K\text{-}{band}\text{:}\mspace{14mu} F_{LO}} = \frac{K - F_{O}}{5}$ ${{for}\mspace{14mu}{the}\mspace{14mu}{Ku}\text{-}{band}\text{:}\mspace{14mu} F_{LO}} = \frac{{Ku} - F_{O}}{3}$ ${{for}\mspace{14mu}{the}\mspace{14mu} X\text{-}{band}\text{:}\mspace{14mu} F_{LO}} = \frac{X - F_{O}}{2}$

This general structure for the F_(LO) signal for each of the radar bands is the same and depicted in FIG. 5 as the line 506. When the FPGA 176 starts the sweep of a particular band, it will control the VCO 140 to start at a frequency 502, sweep at a particular chirp rate r=Δf/Δt, and stop at a frequency 504. The table below assumes the chirp rate for the composite local oscillator to be −3.5 MHz/μs; however, as described above, the VCO 140 is controlled appropriately to produce a sweeping F_(LO) for each band's signal path in order to achieve this composite chirp rate. The table below provides exemplary values for sweeping the four radar bands.

RADAR VCO CHIRP START STOP FREQ SWEEP BAND RATE r FREQ f₁ f₂ TIME Ka $\frac{{- 3.5}\mspace{14mu}{{MHz}/\mu}\; s}{7}$ 5.114 GHz 4.724 GHz ~780 μs K $\frac{{- 3.5}\mspace{14mu}{{MHz}/\mu}\; s}{5}$ 4.797 GHz 4.751 GHz  ~66 μs Ku $\frac{{- 3.5}\mspace{14mu}{{MHz}/\mu}\; s}{3}$ 4.413 GHz 4.367 GHz  ~40 μs X $\frac{{- 3.5}\mspace{14mu}{{MHz}/\mu}\; s}{2}$ 5.138 GHz 5.103 GHz  ~20 μs

Since the sweep signal F_(LO) from the VCO 140 is controlled with high accuracy, determining a timing of the detected pulse within the sweep duration establishes its frequency. In particular, with respect to FIG. 3D, the FPGA 176 determines that a pulse occurs at time T 306 relative to a sweep starting time of a particular sweep of F_(LO) through one of the bands of interest. Such a sweep of F_(LO) is depicted as line 406 in FIG. 4C. In particular, the sweep starts at a time t_(B) 410 and ends at a time t_(E) 412. During that sweep, there is a point t_(P) 414 that is the same distance t_(S) 416 in time from t_(B) 410 as the pulse time T 306 is from the sweep start time t_(B) 410.

As shown in the above table, during the sweep of F_(LO), the sweep begins at a starting frequency f₁ and ends at a stopping frequency f₂. The frequency of F_(LO) at location t_(P) 414 in the sweep is provided by: instantaneous frequency of F _(LO) =f ₁+(t _(S) ×r) This instantaneous value of F_(LO) can then be used to determine the frequency of the radar signal that corresponds to that portion of the sweep of F_(LO).

For example, if a particular sweep of F_(LO) occurs for the X-band to produce an output pulse that is 10 μs after the beginning t_(B) 410 of the sweep, then t_(S) will equal 10 μs. Using the values f₁=5.114 GHz,

$\left. {r = \frac{{- 3.5}\mspace{14mu}{{MHz}/{\mu s}}}{2}} \right),$ and t_(S)=10 μs, will determine that F_(LO)=5.1225 GHz. Using this instantaneous frequency for F_(LO), the relationship of FIG. 2D

$\left( {{e.g.},{F_{LO} = \frac{X - F_{O}}{2}}} \right)$ and F_(O)=280 MHz, will reveal a value of X=10.525 GHz. Thus, the timing T 306 of the detected output pulse provides an indication that the radar signal received at an antenna had a frequency of 10.525 GHz.

As mentioned above, there are four different frequency bands that can be swept through in order to detect possible radar gun signals. In addition to these FCC licensed radar bands, it is advantageous to examine other microwave frequencies to identify nuisance signals leaking from the local oscillators of other radar detectors operating in the vicinity (e.g. 32.55 GHz to 32.65 GHZ and 22.45 GHz to 23.79 GHz). If these nuisance signals can be identified, the false alarms they generate in Ka band may be suppressed. These problems and techniques for their amelioration are taught in U.S. Pat. No. 7,579,976, “SYSTEMS AND METHODS FOR DISCRIMINATING SIGNALS IN A MULTI-BAND DETECTOR” which is herein incorporated by reference in its entirety.

Additionally, in order to determine signal signatures that help differentiate nuisance signals from actual radar gun signals, the four different bands can be swept through multiple times during one complete “sweep” of the radar detector. Thus, the FPGA 176 can control the VCO 140 to sequentially produce appropriate sweep signals (according to the above table) to sweep through each of the four frequency bands. A sweep through all of the four bands can be accomplished in about 1 ms which allows an opportunity to perform multiple sweeps of the four bands within a 16 ms transmission window corresponding to some conventional brief-duration radar signal sources.

For example, FIG. 6 depicts a flowchart of an exemplary sweep process. The order in which a front or a rear antenna is selected (e.g., steps 602, 612) may vary without departing from the scope of the present invention. Similarly, the order in which the four different bands are swept (e.g., steps 604, 606, 608, 610) may vary as well. An example sweep may start, in step 602, by selecting one of the antennas, such as a front antenna. The VCO 140 can then be controlled to sequentially sweep through the X-band (step 604), the Ku-band (step 606), the Ka-band (step 608), and the K-band (step 610) as described in more detail above.

Once the four bands have been swept for the selected antenna, then the other antenna, such as the rear antenna, can be selected in step 612. Similar to the sweep using the front antenna, the four bands are sequentially swept in steps 614, 616, 618, and 620. Once the four bands have been swept for both the front and rear antenna, then a determination is made in step 622 if the sweeps should be repeated. For example, as described earlier, a sweep of all four bands can be accomplished in about 1 ms and, thus, a sweep for both the front and rear antenna can be accomplished in about 2 ms. If a shortest police radar transmission signal lasts about 16 ms, then about 6 or 7 sweeps of all four bands for both antennas can be repeated within that time frame. One of ordinary skill will recognize that the determination in step 622 of how many sweeps to repeat can vary without departing from the scope of the present invention.

Practical design considerations may reduce the number of sweeps that may theoretically be performed during a predetermined time period (e.g., 16 ms) when implementing the sweeping local oscillator. For example, when changing from one sweep rate for a particular band to another sweep rate for another band, the PLL 152 may have a settling time that can be accounted for. One way to account for the settling time is to start the VCO 140, for a particular band sweep, at a F_(LO) frequency higher than the starting frequency f₁ shown in the above table. Following this practice ensures that the PLL loop has settled and the VCO 140 is accurately sweeping when the starting frequency f₁ is reached. For an example explored below, assume the required settling time is 10 μs; during this interval 35 MHz of RF spectrum will be swept.

The DDL SAW filter 170 also has an inherent delay, D_(T), that is accounted for as well. Assume as an example this delay is 4 μs. During this delay 14 MHz of RF spectrum will be swept (for example, in the Ka-band). This delay can be compensated by sweeping 7 MHz before the highest RF frequency for that band and 7 MHz below the lowest RF frequency for that band. Further, to accommodate the PLL settling time mentioned above, an additional 35 MHz can be incorporated above the nominal start of the band, totaling 42 MHz adjustment of the starting point of the RF sweep. Continuing the numerical example, the Ka-band may have a high-end RF frequency of 36.08 GHz and a low-end frequency of 33.35 GHz. To accommodate the delays described above, the sweep generating hardware can be designed to target 36.122 GHz and 33.343 GHz as the start and end points of the RF sweep.

Accounting for the settling time of the PLL 152 and the delay of the filter 170 plus some “programming overhead” that may be associated with the time required to load control settings into the reference generator 156 may result in a complete sweep of both front and rear antennas taking about 3 ms. Even with this longer sweep period, the detector of FIG. 1B can complete about 5 of these full sweeps in a 16 ms time period.

Once all the sweeps have been accomplished, then, in step 624, the detector can provide alarms if a police radar signal was detected and can also ignore any nuisance signals that, even though detected, do not correspond to actual police radar signals.

One known issue with using a mixer in super-heterodyne receivers for combining a desired RF signal and a local oscillator signal to produce an intermediate frequency signal, is that of image signals that interfere with receiving and detecting the desired RF signal. Because a radar detector sweeps a spectrum far larger than the final IF center-frequency, in the presence of a strong signal it is almost inevitable that the detector must contend with a spurious final IF image response. An example will illustrate: Assume a strong radar signal at 35.0 GHz and an equivalent LO at 34.72 GHz, yielding the desired receiver response: 35.0−34.72=0.28 GHz. But the detector's LO will also be swept through 35.28 GHz, yielding 35.0−35.28=−0.28 GHz, i.e. the receiver's IF image. Thus, the strong 35 GHz signal will be seen a second time, but erroneously. Obviously, it is beneficial to design the detector so that this undesired image signal is attenuated sufficiently.

One of ordinary skill will recognize that there are traditional image signal rejection techniques that can be used to filter image signals prior to the final mixer M4 162 (or more generally, before the mixer 54 of FIG. 1A). However, the characteristics of the DDL SAW filter 170 provide additional techniques for rejecting image signals.

Assuming, for example, that the DDL filter has delay length of 4 μs and a characteristic chirp rate of 3.5×10¹² Hz/s, then its bandwidth is approximately (4 μs*3.5 MHz/μs)=14 MHz. If the filter is presented with an up-chirp at the characteristic rate, its resulting output is a narrow pulse lasting a few tens of nanoseconds. Conversely, if presented with a down-chirp, the filter expands rather than compresses the duration of its output. More specifically, if presented with a 3.5×10¹² Hz/s down-chirp (e.g. the “image chirp” produced when the receiver sweeps through the IF image), the filter output will last about 8 μs, twice the filter's delay length, and about 28 MHz of spectrum is swept during this interval.

The result is that the DDL filter inherently enhances the S/N ratio of correctly-chirped signals but attenuates chirp images by dispersing the energy, typically yielding more than 20 dB rejection of image chirps. This phenomenon is depicted in FIG. 7 which overlays a strong desired compressed chirp 710, having a width T_(cc) 702) with the dispersed response 720 arising from reception of an image chirp and having a width of T_(SIC) 704. It should be noted the receiver and DDL filter are linear elements so that image suppression prior to the final mixer and chirp-image suppression inherent in the DDL are independent contributors to image suppression. Additional post-demodulation image attenuation is possible after the demodulating logarithmic amplifier by virtue of the marked difference in pulse 710 width of the desired response (T_(cc)˜70 ns) versus the image pulse width (T_(SIC)˜8 μs). This additional filtering may be implemented with analog techniques and/or digitally within FPGA algorithms. Each of these image suppression mechanisms combines to improve image signal rejection.

FIG. 8A depicts a flowchart of an exemplary process to implement a fixed detection window in accordance with the principles of the present invention. As mentioned earlier, the output from the filter 170 is received by an amplifier 172 that provides input to an analog-to-digital converter 174. The FPGA 176 samples the output of the A/D converter 174 in time periods that can conveniently be referred to as “buckets” such that during a particular bucket, the FPGA 176 determines an input value of the digital signal provided by the A/D converter 174. Buckets can be sequentially numbered so that when the time period for bucket n is finished, the FPGA 176 can begin sampling the input from the A/D converter 174 for bucket n+1. If the time period is known for each bucket and the bucket number is reset at the start of the sweep, the bucket number can be used to calculate the time since the sweep start using the formula t _(s) =n×t _(bucket)

where n is the bucket number and t_(bucket) is the period of one bucket.

Once the time since the sweep start, t_(s), is known, the techniques described above can be used to calculate the instantaneous frequency. Therefore, the bucket can be used to determine the detected frequency during the detection process.

The overall process of FIG. 8A describes one possible detection routine that can be performed by the FPGA 176 during a sweep of one of the four radar frequency bands. In particular, the routine for one band can have a starting step 802 along with an initialization of a detected input value in step 804. Once the routine is started, the input value (to the FPGA 176) from the digital output of the A/D converter 174 is determined by the FPGA 176. More particularly, the FPGA 176, in step 806, calculates a difference between the input value and a noise floor to determine what is referred to as an “Input Error” value. If the “Input Error” value is greater than a predetermined threshold, then a desired signal is likely being detected and control passes to step 810. If the “Input Error” value does not exceed the predetermined threshold, then control passes to step 808.

In step 808, the FPGA 176 can set a variable “maxLevel” to be the value of the currently sampled input from the A/D converter 174. Also, a variable “maxBucket” can be set to the value of the current bucket number. Control can then loop back to step 806, where the input from the A/D converter 174 is once again analyzed to determine if a detected signal is likely present.

Once a likely signal is detected, a timer is started in step 810 and a determination is made in step 812 whether the value in the current bucket (i.e., the input from the A/D converter 174) is greater than the “maxLevel”. As the loop of FIG. 8A is repeated, the value of “maxLevel” will be set to the largest value received as input from the A/D converter 174 and the value of “maxBucket” will be set to the bucket number that corresponds to the largest value received. Thus, if the value in the current bucket is larger than the current value of “maxLevel”, the FPGA 176, in step 814, will replace the value for “maxLevel” with the current A/D input value and replace “maxBucket” with the number of the current bucket.

If the current A/D input is not greater than “maxLevel” or once the “maxLevel” and “maxBucket” values are set, control passes to step 816 where the FPGA 176 determines if 1200 ns has elapsed (e.g., a detection window) since the timer was started in step 810. If no, then control loops back to step 812 to test the current A/D input value. During the time taken to return to step 812, a new bucket value and bucket number may have occurred.

Once the timer has elapsed, according to step 816, the FPGA 176, in step 818, places an alert in a queue to possibly be delivered to the user interface 178. Thus, according to the flowchart of FIG. 8A, the FPGA 176 starts a 1200 ns window once the A/D input exceeds a predetermined threshold and within that 1200 ns window, the FPGA 176 determines a maximum value that occurred and the time period (i.e., bucket) in which it occurred. Accordingly, the alert has both a signal strength and a timing tag associated with it.

One of ordinary skill will appreciate that the example 1200 ns detection window can be changed without departing from the scope of the present invention and may, for example, vary based on the shape of the sinc pulse 304 shown in FIG. 7A. Thus, this detection window can have a duration that captures a desired number of side-lobes 712, 714 and ignores the rest. Also, the test in step 816 can be replaced by a predetermined threshold related to a value of the input from the A/D converter 174. Thus, rather than having a fixed time duration for a detection window, the FPGA 176 could monitor the signal input from the A/D converter 174 until that input signal drops below a predetermined threshold value.

The FPGA 176 repeats the routine of FIG. 8A for each of the four radar bands of interest (and for each of the front and rear antennas). Thus, for each band the FPGA 176 can determine if an alert occurred and also determine for each alert an amplitude of a detected peak signal and when that peak signal occurred. As discussed above with respect to FIG. 4C, the time when the peak signal occurred (e.g., time t_(p) 414 of FIG. 4C) reveals the frequency of that peak signal. Accordingly, the FPGA 176 can construct, for each sweep of the four radar bands, a snapshot of what alerts were detected. Because multiple sweeps of the four bands can occur in a particular time period (e.g., 16 ms), a respective snapshot for each sweep of the four bands can be constructed each time a sweep of the four bands is performed (see FIG. 6, step 622).

The FPGA 176 and/or microprocessor 180 can then compare the different snapshots for each of the sweeps to determine which alerts likely correspond to police radar signals and which alerts likely correspond to nuisance signals that can be ignored. Nuisance signals can include, for example, park-assist systems, automatic cruise control radar systems, and harmonics emitted by nearby radar detectors. U.S. Pat. No. 5,852,417 describes a variety of nuisance signals and techniques for discriminating them from detected signals corresponding to actual police radar and is incorporated herein by reference.

For example, police radar sources typically emit a signal having a fixed fundamental frequency within their respective band of operation. While many other signal sources emit at a fundamental frequency, they may also emit at harmonic frequencies as well. Thus, the FPGA 176 may detect the presence of a signal at a particular frequency, f_(x), within one of the bands. Using the snapshots of the different sweeps through the bands, the FPGA 176 can also determine if signals at multiples of f_(x) (e.g., 2f_(x), 3f_(x), etc.) were also detected. If signals at harmonic frequencies of f_(x) were also detected, then the source of the f_(x) signal is likely not a radar gun and an alert at the frequency f_(x) can be ignored.

Because a snapshot of the approximately 3 GHz of spectrum of the four radar bands can be generated in about 3 ms, a single snapshot can capture the occurrence of a nuisance signal and its harmonics even if that nuisance signal is short-lived. Also, a nuisance signal and its accompanying harmonic signals may typically rise and fall at approximately the same time and, therefore, it is beneficial to be able to sweep through a range of frequencies sufficiently fast enough to detect the occurrence of these multiple signals. Thus, in discriminating signals to identify alerts which likely correspond to actual police radar sources, the FPGA 176 can compare data within a snapshot of the swept four radar bands and can compare data within the multiple snapshots of the four swept radar bands.

Additionally, because it is desirable to detect police radar signals as far from the police radar gun as possible, the signals of interest may be subject to significant multipath fading. In such instances, a signal received by the detector will vary greatly in intensity over a relatively short time period. Thus, when the detector is tuned to receive a particular frequency, the received signal at that frequency may have a negligible amplitude and when the received signal has a detectable amplitude, the detector may no longer be tuned to the appropriate frequency. According to the process described with respect to FIG. 6, the speed at which each sweep of a radar band occurs and the repeated number of sweeps improve the likelihood that even a signal suffering from multipath fading can be more reliably detected because the detector will more likely be tuned to detect the received signal during one or more periods at which the received signal exhibits a detectable amplitude.

Thus, each respective snapshot can be thought of as a virtual image of the signal environment represented by the received electromagnetic signals. Rather than relying on only different sweeps of various bands of the electromagnetic signal environment that can occur at a rate that is relatively large as compared to the duration of some of the detected radar signals, the snapshot, or virtual image, provides information about all the swept bands in a timeframe that is beneficial for detecting short-duration radar signals and/or various nuisance signals.

FIG. 8B depicts a block level diagram of utilizing a dynamically adjusted noise floor in accordance with the principles of the present invention. In particular, a current value output from the A/D converter 174 is provided to a data flip flop (DFF) 830. The output 831 from the DFF 830 is multiplied by 256 and then is provided to an adder 834. The adder 834 subtracts a noise-related value 835 from the output value of the multiplier 832. The output from the adder 834 is passed through a DFF 836 to be one of the input values 846 of a comparator 844. The input value 846 reflects a difference between the output from the multiplier 832 and a noise-related value, and can be referred to as “Input Error” as noted above. The noise-related value 835 can be referred to as a “Noise Floor”.

One exemplary noise-related value 835 is shown in FIG. 8B, in which the “Input Error” is first divided by 8, by block 838, and then summed with a previous value 839 of itself using the adder 852. The value 839 can be divided by 256 by block 842 and may represent a running average of the noise present in the signal received as input from the A/D converter 174.

Returning to the comparator 844, the other input value to the comparator 844 is a predetermined threshold value 848. If the “Input Error” 846 is greater than the predetermined threshold value 848, then a signal found value 852 is output from the comparator 844. This signal found value 852 is equivalent to the test analysis of step 806 in FIG. 8A. The components depicted in FIG. 8B may implement a signal detection technique which relies on a dynamically derived noise floor such that a signal is detected only when an input signal value is greater than the dynamically calculated noise floor by at least a predetermined amount.

In practice, the emitter circuitry of a police radar gun does not instantly turn on at a desired frequency. Instead, due to thermal expansion transients, the typical initial emission from the radar gun is at a higher frequency which ramps down quickly to its desired frequency of operation. This phenomenon can be captured by a radar detector operating in accordance with the principles of the present invention.

In particular, if four combined sweeps through four different radar bands constitute a full sweep cycle for an antenna (e.g., a front antenna), then activation of the emitter circuitry of a radar gun can potentially be detected in each of the four different sweep cycles. FIG. 9 depicts what a detected signal of the activation of a Ka band radar gun may look like; it shows detected signals from four different Ka sweeps on a single time line. During the first sweep 904, a detected signal is determined to occur at t₁ 903 which corresponds to a particular frequency. Similarly, a second signal is detected during the second sweep 907 at t₃ 906, a third signal is detected during a third sweep 910 at t₅ 909, and a fourth signal is detected during a fourth sweep 913 at t₇ 912.

The FPGA 176 and/or microprocessor 180 can include signal analysis routines that recognize the pattern of these four signals as potentially representing the operation of a police radar gun even though the radar gun has yet to emit a radar signal at its desired frequency.

FIG. 10 depicts a block level diagram of components of a portion of radar detector in accordance with the principles of the present invention. The components and their arrangement of FIG. 10 is an example of alternative circuitry that provides functionality similar to that of the circuitry of FIG. 1B. Those components in FIG. 10 that are substantially similar to those of FIG. 1B are labeled with the same element number as in FIG. 1B and a more detailed explanation of these components are provided above with respect to FIG. 1B. One way in which the circuitry of FIG. 10 deviates from that of FIG. 1B is that, although the same number of mixers are employed, there is a separate signal path (i.e., separate mixers 1023, 1035) for each of the K-band and the Ka-band. As a result, leakage of F_(LO) frequencies into downstream IF amplifiers can be suppressed.

In addition to rejecting image frequencies when processing 2F_(LO)±RF products, another advantage of the separate K/Ka mixer design is that rejection of F_(LO)+RF products can be accomplished as well. Unlike the current design of FIG. 10, when the K-band and the Ka-band share a common signal path certain mixer products may cause out-of-band frequencies to be treated as valid radar signals. For example, when a common K/Ka-band signal path is shared, products generated by the K-band mixer that may be rejected by the DDL filter 170 because they are a down-chirp may otherwise appear to be a valid Ka-band signal (and vice versa). The separate signal paths of the design of FIG. 10, allows filter (FLT1) 1011 and filter (FLT2) 1013 to prevent this problem by significantly attenuating the offending frequencies.

The diagram of FIG. 10 shows different switches (e.g., 1001, 1003, 1012, 1027, and 1029) that define the signal path through the radar signal detector at a particular time. These switches may be implemented in a variety of different ways without departing from the scope of the present invention. For example, one or more of the switches may be a transistor having its gate voltage set so as to be transmissive or non-transmissive in a given direction or various switches can be functionally achieved by activating a bias on amplifiers in a desired path and deactivating amplifiers in unwanted paths. For example, the “switches” 1027 and 1029 may not be actual switches. Instead, activating the amplifier 1025 while deactivating the amplifier 1033 is functionally equivalent to the “switch” 1027 being closed while the “switch” 1029 is open. In a similar way, for example, the selective activation of either amplifier 107 or 132 may be utilized to accomplish the functionality of the “switch” 1012. However, to more clearly delineate the different signal paths that may be active during a particular sweep of a particular band, the switches (1001, 1003, 1012, 1027 and 1029) are explicitly depicted in FIG. 10. The microprocessor 180 and/or field programmable gate array (FPGA) 176, or similar control circuitry, can control the switches (or amplifiers) so as to select a current signal path and, thereby, select a particular band of radar signals of interest at a particular time. Thus, in the description below, turning a “switch” on or off is functionally equivalent to activating/deactivating different amplifiers and other components in order to select a particular signal path.

Signals arriving at the antennas (either 102 or 138) are split based on frequency by the diplexer 104 or 136—low frequency signals (X and Ku bands) take one path, and high frequency signals (K and Ka bands) take a different path. K-band and Ka-band frequencies received by the front antenna are amplified by the front first-stage RF amplifier 106. K-band and Ka-band frequencies received by the rear antenna are also amplified by the rear first-stage RF amplifier 134. The first-stage RF amplifiers (106, 134) are switched on or off, along with the 3-way band selector IC switch 1026, to select the signal processing path.

After the received K/Ka frequencies are amplified by the first-stage RF amplifier (106, 134), the front and rear signal paths are combined to form a common path using the combiner 1005. Once combined, K and Ka frequencies are amplified again by a second-stage RF amplifier 1009. After the second-stage RF amplifier 1009, K-band frequencies are separated from Ka-band frequencies by two different frequency-based filters (FLT1) 1011 and (FLT2) 1013. The RF filters (1011, 1013) used to separate the K and Ka bands may also serve as image rejection filters for the succeeding mixing stages. Ka-band frequencies pass to the Ka-band first-stage mixer (M1) 1023, while K-band frequencies pass to the K-band first-stage mixer (M2) 1035. The voltage controlled oscillator (VCO) 140 used for conversion has a fundamental frequency around 5 GHz, which is tripled (around 15 GHz) via the frequency tripler 1019 and then fed to the K/Ka band first-stage mixers 1023, 1035. The second harmonic of the tripler output (around 30 GHz) is used by the first-stage mixers 1023, 1035 to down-convert K and Ka band frequencies and produce the first intermediate frequency (IF1), as explained below with respect to FIG. 11A and FIG. 11B.

In general, though, the VCO 140 provides a local oscillator signal with a frequency F_(LO) of about 5 GHz to the frequency tripler 1019. The output of that tripler 1019, or one of its harmonics, is provided to either the Ka-band frequencies signal path or the K-band frequencies path. For the Ka-band path, the splitter 1031 and the switch 1027 provide the signal to an amplifier 1025 with an output connected to the mixer M1 1023. For the K-band path, the splitter 1031 and the switch 1029 provide the signal to the amplifier 1033 with an output connected to the mixer M2 1035.

X-band and Ku-band frequencies received by the front antenna 102 are amplified by the front first-stage X/Ku amplifier 107. X-band and Ku-band frequencies received by the rear antenna 138 are also amplified by the rear first-stage X/Ku amplifier 132. The first-stage X/Ku amplifiers 107, 132 are switched on or off, along with the 3-way band selector IC switch 1026, to select the signal processing path. After the first-stage amplifier (107 or 132), X and Ku band frequencies received by the antennas (102 or 138) are combined to form a common path.

The output of the front amplifier 107 is joined with the output of the rear amplifier 132 to form a T-junction. The distance from the junction to an output pin of the amplifier 107 is a quarter wavelength of the frequency of interest (e.g., X or Ku band). Likewise, the distance from the junction to an output pin of the amplifier 132 is also a quarter wavelength of the X/Ku band frequency. When the front amplifier 107 is selected, the signal line from the junction to the rear amplifier 132 looks like an RF open circuit; therefore, X and Ku band signals received by the front antenna 102 and amplified by the amplifier 107 pass to the second amplifier 1015 with minimal loss. Conversely, the signal line from the junction to the front amplifier 107 looks like an RF open circuit when the rear amplifier 132 is selected.

Whether from the front amplifier 107 or the rear amplifier 132, X and Ku band frequencies are amplified again by a second-stage X/Ku amplifier 1015. The amplified X/Ku signals then go through an image rejection filter (FLT3) 1017 and pass to the first-stage X/Ku mixer (M3) 1037. The VCO's fundamental output (around 5 GHz) is fed to the X/Ku mixer (M3) 1037. While the X-band frequencies are being processed, the signals input to the mixer (M3) 1037 are mixed with the VCO's fundamental frequency to produce the first intermediate frequency (IF1). However, while Ku-band frequencies are being processed, the signals input to the mixer (M3) 1037 are mixed with the second harmonic of the VCO's fundamental frequency to produce the first intermediate frequency (IF1).

The first intermediate frequencies produced (whether, K, Ka, X or Ku) propagate to the 3-way IC switch 1026 that selects the band being processed. The IF1 selected by the switch 1026 is filtered via filter (FLT4) 1041 and amplified via amplifier 1043 before the second stage mixer (M4) 1047. This filter (FLT4) 1041 may be a low-pass filter designed to suppress leakage of frequencies around F_(LO), harmonic distortion, and intermodulation distortion products from the first-stage mixers 1023, 1035, 1037).

A filter that immediately precedes the second-stage mixer (M4) 1047 is a voltage controlled filter (VCF) 1045 designed to reject image noise. Thus, the filter (FLT5) 1045 is an image rejection filter which prevents undesired frequencies from converting to the desired final intermediate frequency, F_(O). Since F_(O) is 280 MHz, the second-stage mixer 1047 produces this final intermediate frequency when the frequency fed into the mixer 1047 is spaced 280 MHz from the F_(LO) frequency. The frequency spaced 280 MHz above the F_(LO) frequency is the desired RF and the frequency spaced 280 MHz below the F_(LO) frequency is the undesired (image) RF. As the VCO 140 sweeps through F_(LO), the control voltage adjusts the filter's (FLT5) 1045 response accordingly so that the frequency spaced 280 MHz below the F_(LO) frequency is rejected.

The second stage-mixer (M4) 1047 converts IF1 to the final intermediate frequency. The final intermediate frequency (around 280 MHz) is produced by mixing IF1 with the VCO's fundamental frequency. A coupler 1021 is used to provide the output from the VCO 140 to the first-stage mixer (M3) 1037, the second-stage mixer (M4) 1047 and to the PLL 1052.

The final IF from the second-stage mixer (M4) 1047 is amplified via the amplifiers 1049 and 1053 and filtered via the filter (FLT6) 1051 before propagating to the Dispersive Delay Line (DDL) filter 170. The filter (FLT6) 1051 is a bandpass filter that restricts the DDL filter's 170 input signal bandwidth. This filter provides out-of-band rejection by suppressing frequencies that are above and below 280 MHz. When a desired frequency is detected, the DDL produces a narrow pulse, which is processed by the log amplifier and then digitized by the analog-to-digital converter (ADC).

To explain operation of the circuitry in FIG. 10, FIGS. 11A-11D are provided below to isolate each signal path for a respective band of radar signals. In these figures, the various switches, amplifiers, and filters have been omitted for clarity. For each radar band, the VCO 140 produces an appropriately swept F_(LO) that is different from the F_(LO) for the other bands. As depicted in FIG. 11A, a particular spot F_(LO) frequency (e.g., 5.1029 GHz) is shown corresponding to a particular RF signal frequency (e.g., 36 GHz) within a particular radar band (e.g., Ka-band).

Within each of the signal paths depicted in FIGS. 11A-11D there is a respective set of components comprising various mixers and a frequency tripler arranged in a particular way. For each radar band signal path, the respective set of components act together, along with the VCO 140, to achieve a composite local oscillator that functions similar to the linear sweep generator 60 of FIG. 1A. Thus, for each radar band, as F_(LO) is controlled to sweep through its particular range of frequencies at a particular rate, the respective set of components act in concert with one another to function as a composite local oscillator that sweeps through a range of frequencies at a particular rate to produce an intermediate frequency (IF) signal, F_(O), as input to a matched filter. As illustrated below, in order to generate an up-chirp at the input of the DDL filter 170, F_(LO) is swept down in frequency when processing the Ka, Ku, and X bands. On the other hand, F_(LO) is swept up in frequency when processing the K-band.

FIG. 11A depicts major components of the Ka-band signal path from the front antenna 102 (or the rear antenna 138) of the radar signal detector of FIG. 10. The sweep output F_(LO) from the VCO 140 is tripled by a frequency tripler 1019 and the second harmonic (e.g., 30.62 GHz) of this tripled frequency is mixed, at mixer (M1) 1023, with an incoming Ka-band 36 GHz signal. The first intermediate frequency is in the range of 5.38 GHz and is mixed, at mixer (M4) 1047, with the F_(LO) to produce an output signal F_(O) having an intermediate frequency in the range of 280 MHz. This signal is then propagated to the matched filter 170.

As mentioned, the sweep for Ka-band signals may sweep from 36.08 GHz to 33.35 GHz. Using the signal path of FIG. 11A, this sweep can be accomplished when F_(LO) is swept down from 5.1142 GHz to 4.7242 GHz. In other words, when F_(LO) is swept down in frequency, the Ka mixer sweeps from 36.08 GHz to 33.35 GHz.

FIG. 11B depicts major components of the K-band signal path from the front antenna 102 (or the rear antenna 138) of the radar signal detector of FIG. 10. The sweep output F_(LO) (e.g., 4.9060 GHz) is tripled by the frequency tripler 1019 and its second harmonic (e.g., 29.436 GHz) is mixed, at mixer (M2) 1035, with an incoming K-band 24.25 GHz signal. The output of the mixer (M2) 1035 has an intermediate frequency of about 5.186 GHz and is mixed, at mixer (M4) 1047, with F_(LO) to produce an output signal F_(O) having an intermediate frequency in the range of 280 MHz.

Unlike the other three bands, in this example embodiment, the F_(LO) chirp is an up-chirp (i.e., the frequency increases during the sweep). Thus, the sweep for K-band signals may sweep from 24.035 GHz to 24.265 GHz. Using the signal path of FIG. 11B, this sweep can be accomplished when F_(LO) is swept from 4.862 GHz to 4.909 GHz.

FIG. 11C depicts major components of the Ku-band signal path from the front antenna 102 (or the rear antenna 138) of the radar signal detector of FIG. 10. A second harmonic (e.g., 8.8134 GHz) of the sweep output F_(LO) (e.g., 4.4067) is mixed, at mixer (M3) 1037, with an incoming Ku-band 13.5 GHz signal. The output of the mixer (M3) 1037 has an intermediate frequency of about 4.6866 GHz and is mixed, at mixer (M4) 1047, with F_(LO) to produce an output signal F_(O) having an intermediate frequency in the range of 280 MHz.

As mentioned, the sweep for Ku-band signals may sweep from 13.52 GHz to 13.38 GHz. Using the signal path of FIG. 11C, this sweep can be accomplished when F_(LO) is swept from 4.4133 GHz to 4.3667 GHz.

FIG. 11D depicts major components of the X-band signal path from the front antenna 102 (or the rear antenna 138) of the radar signal detector of FIG. 10. The sweep output F_(LO) (e.g., 5.1350 GHz) is mixed, at mixer (M3) 1037, with an incoming X-band 10.55 GHz signal to produce a first intermediate frequency signal in the range of 5.415 GHz. The first intermediate frequency signal is mixed, at mixer (M4) 1047, with F_(LO) to produce an output signal F_(O) having an intermediate frequency in the range of 280 MHz.

As mentioned, the sweep for X-band signals may sweep from 10.56 GHz to 10.49 GHz. Using the signal path of FIG. 11D, this sweep can be accomplished when F_(LO) is swept from 5.14 GHz to 5.105 GHz.

As mentioned above, FIGS. 11A-11D are provided to illustrate an isolated signal path for each respective band of radar signals. In these figures, the various switches, amplifiers, and filters have been omitted for clarity and each of the signal paths happen to relate to either the front antenna 102 or the rear antenna 138 of FIG. 10. Unlike the circuitry of FIG. 1B, the circuitry of FIG. 10 utilizes the same first-stage mixers (e.g., M1, M2, or M3) and second-stage mixer M4, for the signal paths of both the front antenna 102 and rear antenna 138.

FIGS. 11A-11D are similar to FIGS. 2A and 2D in many respects. Using FIG. 11A as an example, the frequency F_(LO) can be characterized as

$F_{LO} = \frac{{Ka} - F_{O}}{7}$ where Ka=36 GHz and F_(O)=280 MHz, for example. As shown in FIG. 11B-11DC, a similar relationship between F_(LO) and a received radar signal band can be determined for the other three bands such as, for example, by:

${{for}\mspace{14mu}{the}\mspace{14mu} K\text{-}{band}\text{:}\mspace{14mu} F_{LO}} = \frac{K - F_{O}}{5}$ ${{for}\mspace{14mu}{the}\mspace{14mu}{Ku}\text{-}{band}\text{:}\mspace{14mu} F_{LO}} = \frac{{Ku} - F_{O}}{3}$ ${{for}\mspace{14mu}{the}\mspace{14mu} X\text{-}{band}\text{:}\mspace{14mu} F_{LO}} = \frac{X - F_{O}}{2}$

This general structure for the F_(LO) signal for each of the radar bands is the same and depicted in FIG. 5 as the line 506. When the FPGA 176 starts the sweep of a particular band, it will control the VCO 140 to start at a frequency 502, sweep at a particular chirp rate r=Δf/Δt, and stop at a frequency 504. The table below assumes the magnitude of the chirp rate for the composite local oscillator to be 3.5 MHz/μs; however, as described above, the VCO 140 is controlled appropriately to produce a sweeping F_(LO) for each band's signal path in order to achieve this composite chirp rate. The table below provides exemplary values for sweeping the four radar bands.

RADAR VCO CHIRP START STOP FREQ SWEEP BAND RATE r FREQ f₁ f₂ TIME Ka $\frac{{- 3.5}\mspace{14mu}{{MHz}/\mu}\; s}{7}$ 5.114 GHz 4.724 GHz ~780 μs K $\frac{{+ 3.5}\mspace{14mu}{{MHz}/\mu}\; s}{5}$ 4.862 GHz 4.909 GHz  ~66 μs Ku $\frac{{- 3.5}\mspace{14mu}{{MHz}/\mu}\; s}{3}$ 4.413 GHz 4.367 GHz  ~40 μs X $\frac{{- 3.5}\mspace{14mu}{{MHz}/\mu}\; s}{2}$ 5.138 GHz 5.103 GHz  ~20 μs

While particular embodiments of the present invention have been illustrated and described, it would be obvious to those skilled in the art that various other changes and modifications can be made without departing from the spirit and scope of the invention. It is therefore intended to cover in the appended claims all such changes and modifications that are within the scope of this invention. 

What is claimed is:
 1. A detector for detecting continuous wave police radar comprising: an antenna configured to receive an input signal, the input signal comprising a continuous wave emission within at least one radar band; a diplexer in communication with the antenna to separate the input signal into a high-band signal having a first range of frequencies and a low-band signal having a second range of frequencies, the second range of frequencies less than the first range of frequencies; a local oscillator configured to sweep through a range of frequencies from a first frequency to a second frequency in a predetermined time period to produce a signal F_(LO); a frequency multiplier in communication with the local oscillator signal to generate a first mixing signal with a frequency that is a positive, integer multiple of F_(LO); a high-band first-stage mixer configured to mix the first mixing signal with the high-band signal to produce a high-band intermediate-frequency signal; a low-band first-stage mixer configured to mix the signal F_(LO) with the low-band signal to produce a low-band intermediate-frequency signal; a switch coupled with the high-band first-stage mixer and the low-band first-stage mixer, configured to select one of the high-band intermediate frequency signal and the low-band-frequency signal as an output intermediate-frequency signal; a second-stage mixer in communication with the switch configured to mix the output intermediate-frequency signal with the signal F_(LO) to generate an output signal F_(O); and a signal analyzer configured to determine whether the input signal received by the antenna includes a police radar signal based on the output signal F_(O).
 2. The detector of claim 1, comprising: a dispersive delay line filter configured to filter the output signal F_(O) to produce a filtered output signal; and wherein the signal analyzer is configured to determine that the input signal received by the antenna includes the police radar signal based on the filtered output signal.
 3. The detector of claim 1, wherein the frequency multiplier comprises a frequency tripler.
 4. The detector of claim 3, wherein the first mixing signal comprises a harmonic of an output of the frequency tripler.
 5. The detector of claim 1, wherein: the high-band signal comprises a first band signal and a second band signal; and the high-band first-stage mixer comprises a first mixer and a second mixer.
 6. The detector of claim 5, comprising: a first high-band filter configured to select the first band signal from the high-band signal; a second high-band filter configured to select the second band signal from the high-band signal; the first mixer in communication with the first high-band filter and configured to mix the first mixing signal with the first band signal to produce a first possible high-band intermediate frequency signal; the second mixer in communication with the second high-band filter and configured to mix the first mixing signal with the second band signal to produce a second possible high-band intermediate frequency signal; and a high-band switch in communication with the first mixer and the second mixer, and configured to select one of the first and second possible high-band intermediate frequency signals as the high-band intermediate-frequency signal.
 7. The detector of claim 1, wherein: the antenna comprises a front antenna and a rear antenna; the diplexer comprises a front diplexer and a rear diplexer; and the input signal comprises a front input signal and a rear input signal.
 8. The detector of claim 7, comprising: the front diplexer in communication with the front antenna to separate the front input signal into a first possible high-band signal having the first range of frequencies and a first possible low-band signal having the second range of frequencies; the rear diplexer in communication with the rear antenna to separate the rear input signal into a second possible high-band signal having the first range of frequencies and a second possible low-band signal having the second range of frequencies; a high-band switch in communication with the front diplexer and the rear diplexer, and configured to select one of the first as second possible high-band signals as the high-band signal; and a low-band switch in communication with the front diplexer and the rear diplexer, and configured to select one of the first and second possible low-band signals as the low-band signal.
 9. The detector of claim 5, wherein: the first band signal comprises the Ka-band, the second band signal comprises the K-band, and the low-band signal comprises at least one of the Ku-band and the X-band.
 10. The detector of claim 5, wherein the first mixing signal has a frequency that is between respective frequencies of the first band signal and the second band signal.
 11. A method for detecting continuous wave police radar comprising: receiving, by an antenna, an input signal, the input signal comprising a continuous wave emission within at least one radar band; separating, by a diplexer in communication with the antenna, the input signal into a high-band signal having a first range of frequencies and a low-band signal having a second range of frequencies, the second range of frequencies less than the first range of frequencies; sweeping, by a local oscillator, through a range of frequencies from a first frequency to a second frequency in a predetermined time period to produce a signal F_(LO); generating, by a frequency multiplier in communication with the local oscillator signal to generate a first mixing signal with a frequency that is a positive, integer multiple of F_(LO); mixing, by a high-band first-stage mixer, the first mixing signal with the high-band signal to produce a high-band intermediate-frequency signal; mixing, by a low-band first-stage mixer, the signal F_(LO) with the low-band signal to produce a low-band intermediate-frequency signal; selecting, by a switch coupled with the high-band first-stage mixer and the low-band first-stage mixer, one of the high-band intermediate frequency signal and the low-band-frequency signal as an output intermediate-frequency signal; mixing, by a second-stage mixer in communication with the switch, the output intermediate-frequency signal with the signal F_(LO) to generate an output signal F_(O); and determining, by a signal analyzer, whether the input signal received by the antenna includes a police radar signal based on the output signal F_(O).
 12. The method of claim 11, comprising: a dispersive delay line filter configured to filter the output signal F_(O) to produce a filtered output signal; and wherein the signal analyzer is configured to determine that the input signal received by the antenna includes the police radar signal based on the filtered output signal.
 13. The method of claim 11, wherein the frequency multiplier comprises a frequency tripler.
 14. The method of claim 13, wherein the first mixing signal comprises a harmonic of an output of the frequency tripler.
 15. The method of claim 11, wherein: the high-band signal comprises a first band signal and a second band signal; and the high-band first-stage mixer comprises a first mixer and a second mixer.
 16. The method of claim 15, comprising: performing one of: a1) selecting, by a first high-band filter, the first band signal from the high-band signal; and a2) mixing, by the first mixer in communication with the first high-band filter, the first mixing signal with the first band signal to produce a first possible high-band intermediate frequency signal; or b1) selecting, by a second high-band filter, the second band signal from the high-band signal; b2) mixing, by the second mixer in communication with the second high-band filter, the first mixing signal with the second band signal to produce a second possible high-band intermediate frequency signal; and selecting, by a high-band switch in communication with the first mixer and the second mixer, one of the first and second possible high-band intermediate frequency signals as the high-band intermediate-frequency signal.
 17. The method of claim 11, wherein: the antenna comprises a front antenna and a rear antenna; the diplexer comprises a front diplexer and a rear diplexer; and the input signal comprises a front input signal and a rear input signal.
 18. The method of claim 17, comprising: separating, by the front diplexer in communication with the front antenna, the front input signal into a first possible high-band signal having the first range of frequencies and a first possible low-band signal having the second range of frequencies; separating, by the rear diplexer in communication with the rear antenna, the rear input signal into a second possible high-band signal having the first range of frequencies and a second possible low-band signal having the second range of frequencies; selecting, by a high-band switch in communication with the front diplexer and the rear diplexer, one of the first as second possible high-band signals as the high-band signal; and selecting, by a low-band switch in communication with the front diplexer and the rear diplexer, one of the first and second possible low-band signals as the low-band signal.
 19. The method of claim 15, wherein: the first band signal comprises the Ka-band, the second band signal comprises the K-band, and the low-band signal comprises at least one of the Ku-band and the X-band.
 20. The method of claim 15, wherein the first mixing signal has a frequency that is between respective frequencies of the first band signal and the second band signal. 